It has long been known that non-linear classes of RF power amplifier operation which switch the RF devices on and off in or near saturation (such as in Class C, D, E, and F amplifiers) are more efficient than linear classes of operation such as Class A or Class AB in which the devices conduct during more than half of the RF cycle and are backed off significantly from compression. However, such non-linear modes of operation are convenient to use only for constant-envelope forms of modulation, such as frequency modulation (FM) and certain forms of phase modulation (PM). In order to apply non-constant envelope modulation methods {such as quadrature amplitude modulation (QAM) or single side band (SSB) or large-carrier AM or multiple-subchannel FM}, a linear transmitter is required.
Prior art solid state linear transmitters have used low-efficiency classes of operation such as Class A or AB in which the conduction angle exceeds 180 degrees, backed off from compression, in order to obtain sufficient linearity to avoid creating undesired spectral components consisting of sums and differences of various modulation frequencies (known as intermodulation distortion, or "IM").
Generally linearity may be broken down into two components: amplitude and phase linearity, as illustrated in FIGS. 1 and 2, respectively. These figures assume a linear class of operation (A or AB) and that the power output is controlled by varying the input signal level. An ideal linear amplifier would have a perfectly linear P-out/P-in characteristic, as shown in the dashed line 402, and phase shift which is unaffected by changes in power level. Since realizable power amplifiers (in any class of operation) suffer significantly from both forms of nonlinearity, as depicted by the solid lines 404, 410, both forms must be corrected. One type of correction uses feedback to perform linearity correction. Another type of correction, predistortion, performs an inverse amplitude and phase correction ahead of the amplifier such that the desired output is achieved. Fixed predistortion utilizes a fixed mapping based on typical measured characteristics, whereas adaptive digital predistortion periodically transmits a training signal which is fed back to a signal processor which adjusts the mapping over time.
A prior-art linear amplifier for amplifying a non-constant envelope signal (in this case large-carrier AM) is shown in FIG. 3; (predistortion or feedback linearity correction is not shown). Note that the envelope of the RF waveform 420 is not constant, and is precisely amplified by the power amplifier 424.
In order to use more efficient classes of RF power device operation while amplifying a non-constant envelope signal (requiring linear amplification), early AM broadcast transmitters utilized Class C plate-modulated final power amplifiers. Modulation was accomplished with large, expensive modulation transformers driven by audio power amplifiers. The modulators were limited in bandwidth (primarily by the transformer), and the audio amplifiers were inefficient, yet it was preferable to shift the dissipation from the final RF device to low-frequency audio power amplifiers in order to obtain higher efficiency in the RF output device(s), which resulted in higher RF output power for a given input power, and reduced power dissipation in the final RF device. Dissipation in the RF devices was (and still is) generally more costly than dissipation in low-frequency modulation devices, due to the higher cost of RF devices and their associated circuitry.
More recently, with the advent of high-efficiency switching regulators and power amplifiers, it has become possible to perform the task of envelope modulation in a way which is both efficient and capable of high bandwidth (with frequency response extending down to DC). FIG. 4 illustrates the use of such techniques to accomplish the required linear amplification using highly non-linear RF amplifiers 444, 446 in the transmitter. Note that the envelope amplitude is constant at the inputs of all the RF amplifier stages, and the supply modulation 452 (typically coupled to the collector or drain of the final stage 494) accomplishes the required amplitude variations (compare with FIG. 3).
For the case of large-carrier AM modulation as shown in FIG. 3 the required modulator bandwidth is approximately equal to the bandwidth of the modulating signals, since in this case there are no RF phase inversions. Unfortunately the modulator bandwidth must be very high for modulation waveforms in which RF phase inversions occur, such as that of 2-tone modulation, as illustrated in FIG. 5. Additionally, it is difficult to pass the phase inversions through the constant-envelope amplifiers, since any band-limiting will cause envelope fluctuations (such envelope fluctuations due to filtering are not shown in the top path of FIG. 5--instead, ideal 180 degree phase inversions are shown). Note the sharp reversals 460 in the switch-mode amplitude modulator output 474 near zero volts, which require much higher modulator bandwidth than the baseband modulating signals.
Another problem with the polar modulation scheme of FIG. 5 is the poor linearity achievable as the supply voltage approaches zero. Due to capacitive coupling within the final devices, the output power does not drop to zero when the supply voltage reaches zero. Also, for supply voltages near zero the phase shift in the final PA stage becomes excessive, due to the fact that the input-output capacitance forms a reactance which is typically larger than the load resistance, causing a bleed-through signal which is shifted 90 degrees from the normal output signal.
Thus, what is needed is an RF power amplifier which eliminates the bleed-through and phase shift effects of the polar modulation scheme as the supply voltage approaches zero, and which also eliminates the need for a modulator with much higher bandwidth than that of the baseband modulation signals, yet retains the high efficiency of the polar modulator.